Low noise switching voltage regulator

ABSTRACT

A low-noise switching voltage regulator for supplying a voltage to a radio frequency (RF) power amplifier is disclosed. In one embodiment, the invention can be conceptualized as a power amplifier supply circuit, comprising a pair of oppositely polarized semiconductor switches, and a data formatter configured to supply a data stream having a voltage transition on at least every other bit to each of the pair of oppositely polarized semiconductor switches.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates generally to controlling radio frequency (RF)transmission power and maximizing battery life in a portablecommunications device. More particularly, the invention relates to a lownoise switching voltage regulator.

2. Related Art

With the increasing availability of efficient, low cost electronicmodules, mobile communication systems are becoming more and morewidespread. For example, there are many variations of communicationschemes in which various frequencies, transmission schemes, modulationtechniques and communication protocols are used to provide two-way voiceand data communications in a handheld, telephone-like communicationhandset. While the different modulation and transmission schemes eachhave advantages and disadvantages, a common goal for all of thesedevices is maximizing the amount of time that a handset can operate on asingle battery charge. This is referred to as maximizing the “talk time”of the handset.

Maximizing the talk time of a portable communication handset istypically the greatest challenge facing the designers of such devices.While many different approaches have been investigated, the singlegreatest improvement in talk time is generally achieved by reducing thepower consumption of the RF PA. Typically, the RF PA consumes thegreatest amount of power in a portable communication handset.

Many techniques have been implemented in an attempt to reduce the powerconsumption of the RF power amplifier. One such technique is the use ofa switching voltage regulator to reduce the supply voltage supplied toan RF power amplifier. By reducing the voltage to the PA it can beoperated closer to the saturation point, and hence the efficiency of thetransmitter can be improved. There are two primary issues that should beresolved before this technique can be practically realized. The firstissue has to do with the output noise or “ripple” of the switchingsupply that will be used to supply the RF PA. Any noise present on thesupply of the RF PA will manifest as noise in the output RF spectrum ofthe RF PA. Most communications standards, such as the global system formobile communications (GSM) have strict specification limits on theamount of noise the system is allowed to add to the transmittedfrequency.

To comply with the power output spectrum standard for GSM, the poweroutput of an RF power amplifier is tested against a specified poweroutput specification, referred to as a power output spectral “mask.” Themask defines an RF spectrum within which the power output of the RFpower amplifier must reside. Out-of-band emission, spurious emission andother factors are tightly controlled. If any portion of the output ofthe RF power amplifier violates the spectrum defined by the mask, theportable communication handset will fail the power spectral mask testand will not be permitted to operate in the communication system. Othercommunication standards, have similar power output mask specifications.

The second issue concerning RF PA saturation demands careful control ofthe RF PA supply signal such that the RF PA power saturation point isabove the desired output power of the system. If these conditions aren'tmet, the power ramp-up and ramp-down required by GSM will allow RF powerto migrate into adjacent channels and could cause the power amplifier tofail the switching transient specification such as those set forth byGSM standard 11.10.

Furthermore, in existing portable communication handsets that use aswitching voltage regulator to supply voltage to an RF power amplifier,the efficiency of the power amplifier is limited due to lack of abilityto reduce the ripple to an acceptable level.

One manner of reducing the ripple is to carefully select components inthe switching voltage regulator to absorb and minimize the ripple.Unfortunately, this typically leads to unacceptably large and costlycomponents.

Therefore, there is a need in the industry to further reduce powerconsumption and increase talk time in a portable communication handset.

SUMMARY

Embodiments of the invention include a low-noise switching voltageregulator for supplying a voltage to a radio frequency (RF) poweramplifier. In one embodiment, the invention can be conceptualized as apower amplifier supply circuit, comprising a pair of oppositelypolarized semiconductor switches, and a data formatter configured tosupply a data stream having a voltage transition on at least every otherbit to each of the pair of oppositely polarized semiconductor switches.

Related systems and methods of operation and computer readable media arealso provided. Other systems, methods, features, and advantages of theinvention will be or become apparent to one with skill in the art uponexamination of the following figures and detailed description. It isintended that all such additional systems, methods, features, andadvantages be included within this description, be within the scope ofthe invention, and be protected by the accompanying claims.

BRIEF DESCRIPTION OF THE FIGURES

The invention can be better understood with reference to the followingfigures. The components within the figures are not necessarily to scale,emphasis instead being placed upon clearly illustrating the principlesof the invention. Moreover, in the figures, like reference numeralsdesignate corresponding parts throughout the different views.

FIG. 1A is a graphical illustration showing the effect of switchingnoise, or ripple, on the output of a conventional switching voltageregulator.

FIG. 1B is a graphical illustration showing the effect of the rippleshown in FIG. 1A on the output spectrum of a portable communicationhandset.

FIG. 2 is a block diagram illustrating a simplified portable transceiver100.

FIG. 3A is a block diagram illustrating a first embodiment of the poweramplifier supply element of FIG. 2.

FIG. 3B is a block diagram illustrating the components within theswitching voltage regulator of FIG. 3A.

FIG. 3C is a block diagram illustrating one exemplary embodiment of thedata formatter of FIG. 3A.

FIG. 4 is a block diagram illustrating an alternative embodiment of thepower amplifier supply element of FIG. 3A.

FIG. 5 is a schematic diagram illustrating the clock data, ordinarytransmit data and Manchester data waveforms.

FIG. 6 is a graphical illustration showing the effect of the Manchesterdata stream supplied to the switching voltage regulator.

FIG. 7 is a block diagram illustrating an alternative embodiment of thepower amplifier supply element of FIGS. 3A and 4.

DETAILED DESCRIPTION

Although described with particular reference to a portable transceiver,the switching voltage regulator can be implemented in any system whereit is desirable to use a switching voltage regulator to control a poweramplifier. The switching voltage regulator, or portions of the controlsystem for the switching voltage regulator, can be implemented insoftware, software, hardware, or a combination of software and hardware.In a preferred embodiment, the switching voltage regulator isimplemented in hardware, as will be described below. The hardwareportion of the invention can be implemented using specialized hardwareelements and logic. Furthermore, the hardware implementation of theswitching voltage regulator can include any or a combination of thefollowing technologies, which are all well known in the art: a discretelogic circuit(s) having logic gates for implementing logic functionsupon data signals, an application specific integrated circuit havingappropriate logic gates, a programmable gate array(s) (PGA), a fieldprogrammable gate array (FPGA), etc.

If control of the switching voltage regulator is implemented insoftware, portions of the switching voltage regulator control softwaremay comprise an ordered listing of executable instructions forimplementing logical functions, and can be embodied in anycomputer-readable medium for use by or in connection with an instructionexecution system, apparatus, or device, such as a computer-based system,processor-containing system, or other system that can fetch theinstructions from the instruction execution system, apparatus, or deviceand execute the instructions.

In the context of this document, a “computer-readable medium” can be anymeans that can contain, store, communicate, propagate, or transport theprogram for use by or in connection with the instruction executionsystem, apparatus, or device. The computer readable medium can be, forexample but not limited to, an electronic, magnetic, optical,electromagnetic, infrared, or semiconductor system, apparatus, device,or propagation medium. More specific examples (a non-exhaustive list) ofthe computer-readable medium would include the following: an electricalconnection (electronic) having one or more wires, a portable computerdiskette (magnetic), a random access memory (RAM), a read-only memory(ROM), an erasable programmable read-only memory (EPROM or Flash memory)(magnetic), an optical fiber (optical), and a portable compact discread-only memory (CDROM) (optical). Note that the computer-readablemedium could even be paper or another suitable medium upon which theprogram is printed, as the program can be electronically captured, viafor instance optical scanning of the paper or other medium, thencompiled, interpreted or otherwise processed in a suitable manner ifnecessary, and then stored in a computer memory.

FIG. 1A is a graphical illustration 10 showing the effect of switchingnoise, or ripple, on the output of a conventional switching voltageregulator. The vertical axis 11 represents voltage and the horizontalaxis 12 represents time. As shown in FIG. 1A a generally triangularwaveform 14 results at the output of a conventional switching voltageregulator. The triangular, or “saw-tooth” waveform is a result of theoperation of the switches within the switching voltage regulator. Theripple as shown in FIG. 1A manifests as an alternating current (AC)signal superimposed on the direct current (DC) output of a conventionalswitching voltage regulator.

When used to control a power amplifier, such as a power amplifierlocated in a portable communication handset, this switching ripple atthe output of a conventional switching voltage regulator manifests asdiscrete energy tones at the switching frequency of the switchingvoltage regulator. These discrete tones are superimposed on the outputof the power amplifier of a portable communication handset. When aportable communication handset is used, for example, in the GSMcommunication system, which has very stringent power outputrequirements, these discrete tones will likely cause the portablecommunication handset to fail the output RF spectrum specification.

FIG. 1B is a graphical illustration 20 showing the effect of the rippleshown in FIG. 1A on the output spectrum of a portable communicationhandset. The vertical axis 21 represents transmit (TX) power and thehorizontal axis 22 represents frequency. The waveform 24 represents thetransmit output of the power amplifier at a fundamental frequency f₁ ofa portable communication handset. The waveform 24 includes a centralportion 25 located at the fundamental frequency f₁ of the transmitchannel and includes energy at the third and fifth order frequencyharmonics in the regions indicated by reference numerals 26 and 27,respectively.

Reference numeral 29 illustrates an exemplary power output mask, whichdefines the transmit power over a given frequency range within which thewaveform 24 must reside to pass the GSM output mask standard. Thisstandard includes limitations on various transmit energy parametersincluding, for example, out-of-band signal propagation and spuriousemission. As shown in FIG. 1B, the output spectrum includes frequencytones, an exemplary one of which is indicated using reference numeral28, that represent the switching ripple produced by a switching voltageregulator. These tones 28, which occur at the frequency and at frequencyharmonics, such as the third and fifth harmonics as shown in FIG. 1B, ofthe switching voltage regulator, are caused by the switching effect ofthe voltage regulator. If the power level of these tones exceed thepower level allowed by the mask 29 for a particular frequency, theportable communication handset will likely fail the output spectral maskspecification. However, even if these tones manifest within the limitsof the mask 29, they may still introduce errors if they are sufficientlyclose to the modulation signal represented by portion 25 of the waveform24.

Accordingly, it is desirable to minimize, or distribute the energy thatis contained in these discrete tones 28 over as broad a frequency rangeas possible, so that these tones do not effect the output of the poweramplifier within the portable communication handset. Essentially, it isdesirable to spread the energy contained in these tones 28 over a broadfrequency range, thus reducing the apparent power in a given bandwidth.

FIG. 2 is a block diagram illustrating a simplified portable transceiver100. Portable transceiver 100 includes speaker 102, display 104,keyboard 106, and microphone 108, all connected to baseband subsystem110. In a particular embodiment, portable transceiver 100 can be, forexample but not limited to, a portable telecommunication handset such asa mobile cellular-type telephone. Speaker 102 and display 104 receivesignals from baseband subsystem 110 via connections 112 and 114,respectively, as known to those skilled in the art. Similarly, keyboard106 and microphone 108 supply signals to baseband subsystem 110 viaconnections 116 and 118, respectively. Baseband subsystem 110 includesmicroprocessor (μP) 120, memory 122, analog circuitry 124, and digitalsignal processor (DSP) 126 in communication via bus 128. Bus 128,although shown as a single bus, may be implemented using multiple bussesconnected as necessary among the subsystems within baseband subsystem110. Microprocessor 120 and memory 122 provide the signal timing,processing and storage functions for portable transceiver 100. Analogcircuitry 124 provides the analog processing functions for the signalswithin baseband subsystem 110. Baseband subsystem 110 provides controlsignals to radio frequency (RF) subsystem 130 via connection 132.Although shown as a single connection 132, the control signals mayoriginate from DSP 126 or from microprocessor 120, and are supplied to avariety of points within RF subsystem 130. It should be noted that, forsimplicity, only the basic components of portable transceiver 100 areillustrated.

Baseband subsystem 110 also includes analog-to-digital converter (ADC)134 and digital-to-analog converters (DACs) 136 and 138. ADC 134, DAC136 and DAC 138 also communicate with microprocessor 120, memory 122,analog circuitry 124 and DSP 126 via bus 128. DAC 136 converts thedigital communication information within baseband subsystem 110 into ananalog signal for transmission to RF subsystem 130 via connection 140.DAC 138 provides a reference voltage power level signal to poweramplifier control element 161 via connection 144. Connection 140, whileshown as two directed arrows, includes the information that is to betransmitted by RF subsystem 130 after conversion from the digital domainto the analog domain.

RF subsystem 130 includes modulator 146, which, after receiving afrequency reference signal, also called a “local oscillator” signal, or“LO,” from synthesizer 148 via connection 150, modulates the receivedanalog information and provides a modulated signal via connection 152 toupconverter 154. The modulated transmit signal may include only phaseinformation, only amplitude information, or both phase and amplitudeinformation, depending on the desired transmit format. Upconverter 154also receives a frequency reference signal from synthesizer 148 viaconnection 156. Synthesizer 148 determines the appropriate frequency towhich upconverter 154 will upconvert the modulated signal on connection152.

Upconverter 154 supplies the modulated signal via connection 158 topower amplifier 160. Power amplifier 160 amplifies the modulated signalon connection 158 to the appropriate power level for transmission viaconnection 162 to antenna 164. Illustratively, switch 166 controlswhether the amplified signal on connection 162 is transferred to antenna164 or whether a received signal from antenna 164 is supplied to filter168. The operation of switch 166 is controlled by a control signal frombaseband subsystem 110 via connection 132. Alternatively, the switch 166may be replaced by a filter (e.g., a diplexer) that allows simultaneouspassage of both transmit signals and receive signals, as known to thosehaving ordinary skill in the art.

The power amplifier receives a supply voltage from the power amplifiersupply element 200 via connection 198. As will be described in furtherdetail below, the power amplifier supply element 200 includes aswitching voltage regulator that provides a low-ripple voltage signal tothe power amplifier 160.

A portion of the amplified transmit signal energy on connection 162 issupplied via connection 170 to power amplifier control element 161. Thepower amplifier control element 161 may form a closed loop output powercontroller utilizing negative feedback to control the output power ofpower amplifier 160 and may also supply an analog power control (APC)signal via connection 172. In an alternative embodiment, as will bedescribed below, the power amplifier supply element 200 may residewithin the power control element.

A signal received by antenna 164 will be directed to receive filter 168.Receive filter 168 will filter the received signal and supply thefiltered signal on connection 174 to low noise amplifier (LNA) 176.Receive filter 168 is a band pass filter, which passes all channels ofthe particular cellular system in which the portable transceiver 100 isoperating. As an example, for a 900 MHz GSM system, receive filter 168would pass all frequencies from 935.2 MHz to 959.8 MHz, covering all 124contiguous channels of 200 kHz each. The purpose of this filter is toreject all frequencies outside the desired region. LNA 176 amplifies thevery weak signal on connection 174 to a level at which downconverter 178can translate the signal from the transmitted frequency back to abaseband frequency. Alternatively, the functionality of LNA 176 anddownconverter 178 can be accomplished using other elements, such as, forexample but not limited to, a low noise block downconverter (LNB).

Downconverter 178 receives a frequency reference signal, also called a“local oscillator” signal, or “LO,” from synthesizer 148, via connection180. The LO signal instructs the downconverter 178 as to the properfrequency to which to downconvert the signal received from LNA 176 viaconnection 182. The downconverted frequency is called the intermediatefrequency or IF. Downconverter 178 sends the downconverted signal viaconnection 184 to channel filter 186, also called the “IF filter.”Channel filter 186 filters the downconverted signal and supplies it viaconnection 188 to amplifier 190. The channel filter 186 selects the onedesired channel and rejects all others. Using the GSM system as anexample, only one of the 124 contiguous channels is actually to bereceived. After all channels are passed by receive filter 168 anddownconverted in frequency by downconverter 178, only the one desiredchannel will appear precisely at the center frequency of channel filter186. The synthesizer 148, by controlling the local oscillator frequencysupplied on connection 180 to downconverter 178, determines the selectedchannel. Amplifier 190 amplifies the received signal and supplies theamplified signal via connection 192 to demodulator 194. Demodulator 194recovers the transmitted analog information and supplies a signalrepresenting this information via connection 196 to ADC 134. ADC 134converts these analog signals to a digital signal at baseband frequencyand transfers the signal via bus 128 to DSP 126 for further processing.

The foregoing description of the receiver components is for exemplarypurposes only. Indeed, other receiver architectures, such as, forexample but not limited to, a super heterodyne receiver, a directconversion receiver, or a sampling receiver, are contemplated to bewithin the scope of the invention.

FIG. 3A is a block diagram 300 illustrating a first embodiment 200 ofthe power amplifier supply element of FIG. 2. The embodiment illustratedin FIG. 3A is an asynchronous implementation of the power amplifiersupply element 200. The power amplifier supply element 200 receives a DCvoltage over connection 226 from, for example, a battery 224. However,it should be mentioned that any type of DC power source can be suppliedvia connection 226 to the power amplifier supply element 200. In apreferred embodiment, the power source 224 is a lithium ion batterysupplying approximately 3.2 to 4.2 volts (V).

The output of the power amplifier supply element 200 is provided onconnection 198 to the power amplifier 160. The power amplifier 160 alsoincludes an input on connection 158 over which data to be transmitted(TX DATA) is modulated onto a high frequency carrier and supplied to thepower amplifier 160. The power amplifier 160 provides an output overconnection 162, which is directed to the antenna 164 (FIG. 2). Toprovide power control, and for exemplary purposes only, a portion of theoutput power of the amplifier 160 on connection 162 is diverted by adirectional coupler 228 via a feedback connection 170 to the poweramplifier control element 161. Other power detection methods, such as DCcurrent sensing, and any other means that provide an accurate indicationof the output power are possible.

The power amplifier control element 161 compares a reference signal onconnection 132 with the feedback signal on connection 170 and develops acontrol signal on connection 172. The power amplifier control element161 then supplies the control signal via connection 172 to the analogpower control (APC) input 230 of the power amplifier 160. In thismanner, the output of the power amplifier is controlled. Those havingordinary skill in the art will understand the operation of the poweramplifier control element 161.

The power amplifier supply element 200 includes an asynchronous switchcontroller 225. The asynchronous switch controller 225 includes a localoscillator 202, which is similar in function to the synthesizer 148 ofFIG. 2, a random data generator 208, and a data formatter 214. Theoutput of the data formatter 214 is supplied to a switching voltageregulator 220 over connections 216 and 217. The local oscillator 202provides a system clock signal over connection 206 to the data formatter214. The local oscillator 202 also provides a local oscillator (LO)reference signal over connection 204 to the random data generator 208.The random data generator 208 uses the LO input over connection 204 togenerate a random data stream over connection 212. The random datastream on connection 212 is supplied to the data formatter 214 alongwith the clock signal from the local oscillator 202 over connection 206.

The implementation shown in FIG. 3A is considered “asynchronous” becausethe data generated by the random data generator 208 and supplied to thedata formatter 214 over connection 212 bears no relation to the transmitdata on connection 158 supplied to the power amplifier 160.

The data formatter 214 preferably provides a Manchester-type coded datastream to the switching voltage regulator 220 via connections 216 and217. Manchester coding provides a voltage transition on every bit in thedata stream. In Manchester coding, a binary “1” is represented by apulse that has a positive voltage during the first half of the bitduration and a negative voltage during the second half of the bitduration. A binary “0” is represented by a pulse that is negative duringthe first half of the bit duration and positive during the second halfof the bit duration. The negative or positive mid-bit transitionindicates a binary 1 or binary 0, respectively. Thus, a Manchester codeis classified as an instantaneous transition code in that it has nomemory.

The random data on connection 212 is converted into a Manchester datastream by the data formatter 214 and supplied to the switching voltageregulator 220 via connections 216 and 217. However, it should be notedthat a Manchester code is only a preferred embodiment. Any other codingscheme can be used, so long as a long string of 1's or 0's withoutvoltage transitions is avoided. Importantly, the signal supplied to theswitching voltage regulator 220 on connections 216 and 217 shouldinclude a sufficient number of voltage transitions so that neither oneof the transistor switches within the switching voltage regulator 220(to be described below), will remain in one state for more that a fewbits. If this happens, the output voltage would either increase ordecrease outside of the acceptable voltage range used by the poweramplifier to meet the spectral mask specification.

By supplying a Manchester data stream to the switching voltage regulator220 on connections 216 and 217, the switching voltage ripple (see FIG.1A) or its frequency distribution (see FIG. 1B, reference numeral 28)illustrated for a non-Manchester case, at the output of the switchingvoltage regulator 220 on connection 198 is distributed over a broadfrequency spectrum. This is so because the output spectrum of Manchestercoding contains no energy at DC and very low energy at low frequenciesas will be described below with respect to FIG. 6.

FIG. 3B is a block diagram illustrating the components within theswitching voltage regulator 220 of FIG. 3A. The switching voltageregulator 220 includes a P-type field effect transistor (FET) 232 and anN-type FET 234. The gate terminal of the FET 232 is coupled to theconnection 216 and the gate terminal of the FET 234 is coupled toconnection 217. The source terminal of FET 232 is coupled to the DCvoltage source on connection 226 while the drain terminal of FET 232 iscoupled to the drain terminal of FET 234 on connection 236 and to aninductor 238. The source terminal of FET 234 is coupled to ground 246via connection 244. The FETs 232 and 234 are activated based on thepolarity of the signal received on connections 216 and 217,respectively. When FET 232 is switched on, FET 234 is switched off. TheManchester data stream on connections 216 and 217 ensure that the FETs232 and 234 are continuously switching. The two FETS 232 and 234 arecontrolled by separate signals via connections 216 and 217 to avoid bothFETS being switched on at the same time. Similarly, the FETS 232 and 234are controlled so that they are not both switched off for more than afew microseconds (μs). To accomplish this, two non-overlapping signalsare supplied by the data formatter 214 via connections 216 and 217, aswill be discussed below.

The output of the FETs 232 and 234 on connection 236 passes through theinductor 238, which is also coupled in parallel to a capacitor 242. Theoutput of the switching voltage regulator 220 is supplied via connection198 to the power amplifier 160.

FIG. 3C is a block diagram illustrating one exemplary embodiment of thedata formatter 214. In the embodiment shown in FIG. 3C, the dataformatter 214 comprises an exclusive (XOR) gate 252 having inputs 206and 212. A clock signal is received from the local oscillator 202 onconnection 206 and the data from the random data generator 208 issupplied to the XOR gate 252 on connection 212. The output of the XORgate 252 on connection 254 is supplied to a gate driver 256. The gatedriver 256 develops output signals on connections 216 and 217. Thesignal on connections 216 and 217 is the Manchester data stream asdescribed above. The outputs on connections 216 and 217 arenon-overlapping signals and are opposite in phase such that the FETs 232and 234 are not switched on simultaneously, and are not both switchedoff for more than a few microseconds.

FIG. 4 is a block diagram 400 illustrating an alternative embodiment 250of the power amplifier supply element 200 of FIG. 3A. The poweramplifier supply element 250 includes a synchronous switch controller235, which includes the data formatter 214. The output of the dataformatter 214 is supplied to the switching voltage regulator 220 overconnections 216 and 217. However, the data supplied to the dataformatter 214 is also the same data used to modulate the high frequencycarrier supplied to the power amplifier 160 via connection 158.

Because the transmit data is also used as input to the data formatter,the Manchester data stream on connections 216 and 217 will introduce avoltage ripple that is coherent and coincident with the transmitteddata. In this manner, the difference between the spectral data andspectral noise due to the random voltage ripple is harder to detect andtherefore more tolerable by the system. A clock signal is also suppliedto the data formatter 214 via connection 254. The remaining componentsof FIG. 4 operate the same as those described above with respect to FIG.3A.

FIG. 5 is a schematic diagram 500 illustrating the clock data, ordinarytransmit data and Manchester data waveforms. The clock waveform 502includes a pulse on every bit cycle. The transmit data waveform 504 isessentially a random data waveform that can include a series of logicones (1's) and logic zeros (0's) in any random order. The Manchesterdata stream 510 is preferred in this embodiment of the invention,because it includes a voltage transition on every bit cycle. While thetransmit data waveform 504 can be relatively random in nature, there aremany instances when a long series of logic 1's or logic 0's may comprisethe transmit data waveform 504. This situation adds to the undesirableswitching ripple because it allows one of the FETs in the switchingvoltage regulator 220 to remain switched on and the other FET to remainswitched off for a period of time exceeding a few microseconds.

FIG. 6 is a graphical illustration 600 showing the effect of theManchester data stream supplied to the switching voltage regulator 220.The vertical axis 602 represents energy while the horizontal axis 604represents frequency. The transmit data stream is represented usingcurve 606 while the Manchester data stream is represented using curve608. Manchester coding distributes the corrupting energy caused by theswitching ripple over the transmit signal band and away from the DClevel. By properly selecting the switching frequency, it is possible todistribute the switching ripple away from the transmit modulation(represented as at a maximum at DC in FIG. 6), but close enough so thatit rolls off outside of the transmit frequency range. In this manner,the transmit waveform can fall within the transmit mask as shown abovein FIG. 1B. By randomizing the data before driving the switching voltageregulator, as described above with respect to FIGS. 3A and 4, theswitching ripple caused by the switching effect of the FETs within theswitching voltage regulator 220 is distributed over a wide frequencyrange, and preferably, away from the transmit data waveform.

As shown in FIG. 6, the transmit data waveform 606 includes portion 616which has its maximum energy level at DC. This energy falls to a minimumat a frequency represented at 1/T, where T is equal to one cycle. TheManchester data waveform 608 has zero energy at DC, and has a smoothramp-up of power, illustrated by portion 610, to a maximum pointindicated by reference numeral 612. This maximum point 612 is distantfrom the point (DC) at which the transmit data is at a maximum. This isbeneficial because the Manchester data spreads the switching ripple overthe frequency range from DC to 2/T and because most of the energy of thetransmit data stream is now located where the switching ripple pollutesthe least. In other words, at DC there is no switching ripple energy.Therefore, when the switching voltage regulator 220 switches data, thereis no switching energy present at DC.

Furthermore, because Manchester data provides an energy transition(i.e., a voltage transition) on every bit, the power amplifier issmoothly controlled. Accordingly, as shown by curve 608, and by curvedportion 614, by Manchester coding the randomized data supplied to theswitching voltage regulator, the total switching ripple energy has amuch lower level than it would if the transmit data alone were used todrive the switching voltage regulator.

FIG. 7 is a block diagram 700 illustrating an alternative embodiment ofthe power amplifier supply element of FIGS. 3A and 4. In the embodimentshown in FIG. 7, the switching voltage regulator 720 is located in apower control loop associated with the power amplifier 160. A referencesignal is supplied over connection 132 to a saturation detection andcompensation element 730 and to a saturation protected power amplifiercontrol (PAC) element 706. The power amplifier control element 706generates a power amplifier control signal on connection 172 andgenerates an offset signal on connection 712. The offset signal onconnection 712 is used by the saturation detection and compensationelement 730 to adjust the output voltage of the switching voltageregulator 720 via a control signal over connection 710.

The data to be transmitted is supplied to the power amplifier 160 viaconnection 708. The output of the power amplifier 160 is supplied onconnection 162 to the antenna 164 (FIG. 2). A directional coupler 228supplies power from the output of the power amplifier 160 via connection170 to the power amplifier control (PAC) element 706. The output of thedirectional coupler 228 on connection 170 is supplied to a rectifier702. The rectifier 702 transforms the radio frequency (RF) signal onconnection 170 into a DC feedback signal that is supplied via connection704 to the power amplifier control element 702.

The power amplifier control element 706 includes either an active orpassive low pass loop filter. The power amplifier control signal outputof the power control element 706 is supplied on connection 172 to theAPC input 230 of the power amplifier 160 so that the output of the poweramplifier 160 is controlled according to the signal provided by thepower control element 706. Under normal circumstances, when the poweramplifier 160 is not saturated, the output power is proportional to thecontrol signal on connection 132.

However, it is possible that the power amplifier 160 can becomesaturated. As the output power of the amplifier 160 increases, theamplifier 160 may enter saturation, in which case any increase in APC(input 230 to the PA 160) does not cause the output power to increase.The ideal output power that maximizes the efficiency of the poweramplifier 160 depends on a number of factors. For example, these factorsinclude supply voltage, the power amplifier input and output matchingnetworks (not shown), the power amplifier design, and the temperature atwhich the power amplifier is operating. Normally, the matching networks(not shown) are embedded in the power amplifier module. However,lowering the supply voltage that is supplied to the power amplifier 160can force the power amplifier 160 into saturation when operating nearmaximum power for the handset.

The power control element 706 also supplies an offset signal overconnection 712 to the saturation detection and compensation element 730.The offset signal on connection 712 is combined with the referencesignal on connection 132 by the saturation detection and compensationelement 730. The saturation detection and compensation element 730provides a control signal to the switching voltage regulator 720 overconnection 710 and can supply a signal to the power amplifier controlelement 706 via connection 712, which can be a bidirectional connection.The output of the saturation detection and compensation element 730 isdetermined by the level of the offset signal on connection 712.

The control signal supplied by the saturation detection and compensationelement 730 on connection 710 determines the output voltage of theswitching voltage regulator 720 on connection 714. The output voltage ofthe switching voltage regulator 720 is adjusted such that the poweramplifier 160 will be operated at such level where the system efficiencywill be maximized.

When implemented outside the power control loop, as shown in FIGS. 3Aand 4, the switching voltage regulator 220 generally providessufficiently high supply voltage to the power amplifier 160 so that thepower amplifier 160 will not enter into saturation. This implies twoconditions. First, the overall system efficiency is not as high as itcould be because it is always desirable to allow the power amplifiermore supply voltage than it needs. This provides a safety margin fordevice variations, variations in temperature, etc. Second, in theimplementation shown above in FIGS. 3A and 4, the switching voltageregulator 220 does not require an analog control signal to determine thevoltage to which its output should regulate.

While various embodiments of the invention have been described, it willbe apparent to those of ordinary skill in the art that many moreembodiments and implementations are possible that are within the scopeof this invention. Accordingly, the invention is not to be restrictedexcept in light of the attached claims and their equivalents.

1. A method for supplying voltage to a power amplifier in a portablecommunication handset, comprising: providing a switching voltageregulator, the switching voltage regulator configured to provide adirect current (DC) output; and providing a data stream to the switchingvoltage regulator, wherein the data stream comprises sufficient bittransitions to spread any alternating current (AC) voltage on the DCoutput over a bandwidth occupied by data to be transmitted by theportable communication handset.
 2. The method of claim 1, furthercomprising converting the data stream to a Manchester line code.
 3. Themethod of claim 2, wherein the data stream is random.
 4. The method ofclaim 2, wherein the data stream is a transmit data stream.
 5. Themethod of claim 1, wherein the switching voltage regulator is located ina power amplifier control loop.
 6. The method of claim 3, wherein theswitching voltage regulator comprises a P-type field effect transistor(FET) and an N-type FET.
 7. The method of claim 2, wherein theManchester line code distributes switching ripple caused by theswitching voltage regulator to a portion of the frequency spectrumhaving minimal transmit energy.